Ultra-wideband communication system

ABSTRACT

In an ultra-wideband (“UWB”) communication system, methods are disclosed for transmitting packets in multiple portions, each having a different pulse repetition frequency (“PRF”). Methods are also disclosed for transmitting packets dis-continuously.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is related to the following:

-   -   1. Provisional Application Ser. No. 62/639,022, filed 6 Mar.        2018 (“First Parent Provisional”);    -   2. Provisional Application Ser. No. 62/667,909, filed 7 May 2018        (“Second Parent Provisional”); and    -   3. Provisional Application Ser. No. 62/695,140, filed 8 Jul.        2018 (“Third Parent Provisional”).

This application claims priority to the First, Second and Third ParentProvisionals, and hereby claims benefit of the filing dates thereofpursuant to 37 CFR § 1.78(a)(4).

The subject matter of the First, Second and Third Parent Provisionals,each in its entirety, is expressly incorporated herein by reference.

BACKGROUND OF THE INVENTION 1. Field of the Invention

The present invention relates generally to wireless communicationsystems, and, in particular, to a wireless communication system havingimproved performance.

2. Description of the Related Art

In general, in the descriptions that follow, we will italicize the firstoccurrence of each special term of art which should be familiar to thoseskilled in the art of ultra-wideband (“UWB”) communication systems. Inaddition, when we first introduce a term that we believe to be new orthat we will use in a context that we believe to be new, we will boldthe term and provide the definition that we intend to apply to thatterm. In addition, throughout this description, we will sometimes usethe terms assert and negate when referring to the rendering of a signal,signal flag, status bit, or similar apparatus into its logically true orlogically false state, respectively, and the term toggle to indicate thelogical inversion of a signal from one logical state to the other.Alternatively, we may refer to the mutually exclusive Boolean states aslogic_0 and logic_1. Of course, as is well known, consistent systemoperation can be obtained by reversing the logic sense of all suchsignals, such that signals described herein as logically true becomelogically false and vice versa. Furthermore, it is of no relevance insuch systems which specific voltage levels are selected to representeach of the logic states.

By way of example, in an ultra-wideband (“UWB”) communication system, aseries of special processing steps are performed by a UWB transmitter toprepare payload data for transmission via a packet-based UWB channel.Upon reception, a corresponding series of reversing steps are performedby a UWB receiver to recover the data payload. Details of both series ofprocessing steps are fully described in IEEE Standards 802.15.4(“802.15.4”) and 802.15.4a (“802.15.4a”) (“Standards”), copies of whichare submitted herewith and which are expressly incorporated herein intheir entirety by reference. As is known, these Standards describerequired functions of both the transmit (“Tx”) and receive (“Rx”)portions of the system, but specify implementation details only of thetransmit portion of the system, leaving to implementers the choice ofhow to implement the receive portion.

One or more of us have developed certain improvements for use in UWBcommunication systems, which improvements are fully described in thefollowing pending applications or issued patents, all of which areexpressly incorporated herein in their entirety:

“A Method and Apparatus for Transmitting and Receiving ConvolutionallyCoded Data”, U.S. Pat. No. 7,636,397, issued 22 Dec. 2009;

“A Method and Apparatus for Generating Codewords”, U.S. Pat. No.7,787,544, issued 31 Jul. 2010;

“A Method and Apparatus for Transmitting and Receiving ConvolutionallyCoded Data”, U.S. Pat. No. 8,358,709, issued 22 Jan. 2013; and

“Receiver for Use in an Ultra-Wideband Communication System”, U.S. Pat.No. 8,437,432, issued 7 May 2013;

“Convolution Code for Use in a Communication System”, U.S. Pat. No.8,677,224, issued 18 Mar. 2014;

“Adaptive Ternary A/D Converter for Use in an Ultra-WidebandCommunication System”, U.S. Pat. No. 8,436,758, issued 7 May 2013;

“Receiver for Use in an Ultra-Wideband Communication System”, U.S. Pat.No. 8,760,334, issued 24 Jun. 2014;

“Receiver for Use in an Ultra-Wideband Communication System”, U.S. Pat.No. 9,054,790, issued 9 Jun. 2015;

“Adaptive Ternary A/D Converter for Use in an Ultra-WidebandCommunication System”, U.S. Pat. No. 9,325,338, issued 26 Apr. 2016; and

“Secure Channel Sounding”, PCT Application EP2017/052564, filed 6 Feb.2017.

In conformance with the Standards, a UWB communication system may beadapted to implement an embodiment of a known 27 Mbps modulation schema.In accordance with this schema, the highest data rate currently definedis 6.8 Mbps at a pulse repetition frequency (“PRF”) of 64 MHz. We submitthat it is both possible and desirable to allow the PRF to vary within apacket.

Even if a typical UWB communication system is adapted to operate at aHigh Rate Pulse (“HRP”), packet transmission is continuous: preamble,SFD, data, plus, maybe, cipher—all concatenated together in a continuoustransmission. In general, this makes it easier to acquire and maintaincarrier synchronization. However, despite causing implementationdifficulties in some implementations of the receiver, we submit thathaving dis-continuous packets will offer advantages.

We submit that what is needed is an improved method and apparatus foruse in the receiver of a wireless communication system to transmitpackets at variable PRF. Further, we submit that such variable PRFpackets be transmitted dis-continuously. In particular, we submit thatsuch a method and apparatus should provide performance generallycomparable to the best prior art techniques, but allow packets to betransmitted dis-continuously.

BRIEF SUMMARY OF THE INVENTION

In accordance with a preferred embodiment of our invention, we provide amethod for use in a wireless communication system for transmitting apacket comprising first and second portions. In particular, the methodcomprises configuring a transmitter facility of the system to performthe steps of: transmitting the first portion of the packet at a selectedfirst pulse repetition frequency (“PRF”); and transmitting the secondportion of the packet at a selected second PRF different from the firstPRF. Further, the method comprises configuring the system to perform thestep of transmitting the packet dis-continuously.

In one further embodiment, a wireless communication system is configuredto perform our method for transmitting dis-continuous packets.

The methods of our invention may be embodied in computer readable codeon a suitable non-transitory computer readable medium such that when aprocessor executes the computer readable code, the processor executesthe respective method.

The methods of our invention may be embodied in non-transitory computerreadable code on a suitable computer readable medium such that when aprocessor executes the computer readable code, the processor executesthe respective method.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

Our invention may be more fully understood by a description of certainpreferred embodiments in conjunction with the attached drawings inwhich:

FIG. 1 illustrates, in block diagram form, one embodiment of a receiveradapted for use in a UWB communication system, the receiver comprisingboth transmission and reception facilities;

FIG. 2 illustrates, in block diagram form, one embodiment of a receiverfacility adapted to practice our invention;

FIG. 3 illustrates, in linear time form, a continuous packettransmission sequence having a bursty preamble;

FIG. 4 illustrates, in linear time form, a continuous packettransmission sequence having a concentrated preamble;

FIG. 5 illustrates, in linear time form, a first possible dis-continuouspacket transmission sequence;

FIG. 6 illustrates, in linear time form, a second possibledis-continuous packet transmission sequence;

FIG. 7 illustrates, in linear time form, a third possible dis-continuouspacket transmission sequence;

FIG. 8 illustrates, in linear time form, a fourth possibledis-continuous packet transmission sequence;

FIG. 9 illustrates, in linear time form, a fifth possible dis-continuouspacket transmission sequence;

FIG. 10 illustrates, in linear time form, a sixth possibledis-continuous packet transmission sequence;

FIG. 11 illustrates, in linear time form, a seventh possibledis-continuous packet transmission sequence;

FIG. 12 illustrates, in linear time form, an eighth possibledis-continuous packet transmission sequence;

FIG. 13 illustrates, in linear time form, a ninth possibledis-continuous packet transmission sequence;

FIG. 14 illustrates, in block diagram form, one embodiment of aconvolutional encoder adapted to practice our invention; and

FIG. 15 illustrates, in graphical diagram form, one example of a chipsequence developed in accordance with our invention.

In the drawings, similar elements will be similarly numbered wheneverpossible. However, this practice is simply for convenience of referenceand to avoid unnecessary proliferation of numbers, and is not intendedto imply or suggest that our invention requires identity in eitherfunction or structure in the several embodiments.

DETAILED DESCRIPTION OF THE INVENTION

Shown by way of example in FIG. 1 is one embodiment of a receiver 10adapted for use in a UWB communication system, the receiver 10comprising both a transmit facility 12 and a receive facility 14. Shownby way of example in FIG. 2 is one embodiment of a receive facility 14adapted to practice our invention. Complete details relating to theconstruction and methods of operation of receiver 10, and the transmitand receive facilities 12-14, may be found in one or more of the patentsset forth above.

In our Third Parent Provisional, we have disclosed several methods forvarying the PRF of different components of a Standard UWB packet. OnSlide 11, we introduce the possibility that the Standards support a 27Mbps data rate at the nominal PRF of 64 MHz wherein, in a first variant,V1, each burst consists of 2 pulses with a 2 ns spacing there between.However, we noted in Slide 12 that this V1 is less than desirable due toa high spectral peak to average ratio (“SPAR”). We therefore proposed inSlide 13 a second variant, V2, in which the pulses per burst isincreased from 2 to 8. Based in part on simulations that we haveperformed, we noted that V2 predicts several important advantages:

-   -   6 dB better performance that of V1;    -   Same range of V1 over the Standard 6.8 Mbps mode (assuming peak        antenna voltage limited to ˜0.7 v; and    -   Power consumption for Tx and Rx data portion of the frame is ¼        that of the Standard 6.8 Mbps mode.

In our First Parent Provisional, we have disclosed the scope and resultsof our simulations that form the basis of this invention. Let us nowsummarize those simulation studies with reference to our First ParentProvisional.

In accordance with our Compressed Modulation Schema (“CMS”), the numberof transmitted chips per input bit and, hence, the number of transmittedchips per transmitted symbol are equal to the logic_1s in the currentlyhighest data rate specified in the Standards, i.e., 6.8 Mbps at a 64 MHzPRF. However, in our CMS, the data rate is four (4) times the highestStandard data rate. Further, in accordance with our CMS, both ofexisting concatenated error correction codes, i.e., Reed-Solomon (“RS”)and convolutional, are preserved and unmodified. In other words, botherror correction coding and decoding schemes are unmodified. What ismodified is the way the convolutional-encoded bits are spread in the Txonto transmitted bursts and, hence, de-spread in the Rx.

In FIG. 14, we have illustrated a convolutional encoder 16 constructedin accordance with the Standards. For the k-th input bit (b^((k))),encoder 16 outputs two bits: a systematic bit (g₀ ^((k))); and a paritybit (g₁ ^((k))). In the IEEE 802.15.4a BPPM-BPSK hybrid modulationscheme the burst position is decided by g₀k) and it is multiplied by abipolar version of g₁ ^((k)). Hop position and scrambling sequence aregenerated by a Standard scrambling m-sequence shift register generator.

In accordance with our CMS, g₁ ^((k)) still multiplies the burst.Furthermore, scrambling sequence is generated in the same way. However,there is no hopping and position modulation, but, rather, the 0-thposition is always used. Now, g₀ ^((k)) decides which of two possiblemutually orthogonal carrier sequences will be used:

$\begin{matrix}{s^{(k)} = \left\{ \begin{matrix}{{s_{0} = \left\lbrack {+ {,\ {+ {,\ {+ {,\ {+ {,\ {+ {,\ {+ {,\ {+ {,\  +}}}}}}}}}}}}}} \right\rbrack},\ {g_{0}^{(k)} = 0},} \\{{s_{1} = \left\lbrack {- {,{- {,{- {,{- {,{+ {,\ {+ {,\ {+ {,\  +}}}}}}}}}}}}}} \right\rbrack},\ {g_{0}^{(k)} = {1.}}}\end{matrix} \right.} & {{Eq}.\mspace{14mu} 1}\end{matrix}$

s^((k)) is then multiplied by the bipolar version of g₁ ^((k)) to getv^((k)):

v ^((k))=(−1)^(g) ¹ ^((k)) s ^((k))  Eq. 2:

v^((k)) is then scrambled by the scrambling sequence and transmitted.

This can be understood more clearly by the following parallel. In theBPPM-BPSK hybrid, bit g₀ ^((k)) places the unscrambled ‘all ones’ burstin two possible positions, each mutually orthogonal in time. In our CMS,g₀ ^((k)) alters the burst itself to use one of two possible unscrambledsequences, each mutually orthogonal in the code space. Notice that thesequence orthogonality is preserved after scrambling. Furthermore, anytwo binary (±1) orthogonal sequences can be used instead of the abovetwo sequences in Eq. 1, and they would provide the same Euclidiandistances between respective constellation points; we have selectedthese examples for simplicity. Note that the length of the sequencesused can also change, e.g., for changing data rate, so long as theorthogonality is preserved.

Our CMS develops symbol intervals of 32 ns duration, each comprising 16chips. The first half of the symbol interval, i.e., comprising 8 chips,is occupied by the scrambled version of v^((k)), whereas the second halfof the symbol interval, also comprising 8 chips, represents a guardinterval. FIG. 15 depicts one example of the chip sequence generated inour Matlab testbench. In this embodiment, both the physical header(“PHR”) and the physical layer (“PHY”) service data unit (“PSDU”) usethe same compressed modulation format.

After channel match filter (“CMF”), rotation, resampling anddescrambling at the chip rate, the receiver 14 will have an estimate ofv^((k)), denoted {circumflex over (v)}^((k)). In order to calculatemetrics for the Viterbi decoding, denoted {circumflex over (p)}₀ ^((k))and {circumflex over (p)}₁ ^((k)), the receiver 14 should project{circumflex over (v)}^((k)) onto sequences s₀ and s₁, respectively:

{circumflex over (p)} ₀ ^((k)) ={circumflex over (v)} ^((k)) s ₀^(T),  Eq. 3a:

={circumflex over (v)} ^((k)) s ₁ ^(T)  Eq. 3b:

You should note that the Viterbi metrics {circumflex over (p)}₀ ^((k))and {circumflex over (p)}₁ ^((k)) are analogous to the metrics atpositions zero (0) and one (1), respectively, of the BPPM-BPSK hybridmodulation. Hence, they are used in place of these metrics as the inputof the Viterbi decoder, carrier loop phasor, etc. In our Matlab code,this is done as follows:

burst=this.resampleOutput(burstOffset+(1: this.nSamplesPerBurst));dscrmbld=burst(1:this.burstDeciRate:end).*(1−2*this.spreadingSeq);sL=sum(dscrmbld(1: this.chipsPerBurst/2));sU=sum(dscrmbld(this.chipsPerBurst/2+1: this.chipsPerBurst));this.despreadOntime0=sL+sU;this.despreadOntime1=−sL+sU;The above Matlab code snippet displays one more important feature: both{circumflex over (p)}₀ ^((k)) and {circumflex over (p)}₁ ^((k)) can becalculated from the same two sums of the descrambled chips, wherein thefirst is the sum of the lower four chips and the second is the sum ofthe upper four chips. This suggests that there is no need to implementtwo descramblers in the hardware, since a small modification of theexisting one probably would suffice.

For the compressed data mode, a carrier loop sampling period of 1024 ns,already used for all implemented data rates, has been preserved. Sincethe symbol period is now equal to 32 ns, this mode uses 1024/32=32smooth steps of the carrier loop for the rotation of the symbols'samples.

By way of completeness, we have provided in the Second ParentProvisional the simulation parameters and performance results of thestudy we performed on our CMS as disclosed herein. As can be seen, thesensitivity of our 27 Mbps compressed data mode is considerably affectedby the 27 Mbps PHR errors. It is known, however, that the PHR is weaklyerror-protected by the SECDED code. This weak PHR protection does notaffect so much the 6.8 Mbps mode sensitivity, since PHR is transmittedat the 8 times lower data rate of 850 kbps, hence, each symbol has 8times (9 dB) higher energy than 6.8 Mbps PSDU symbols. On the otherhand, the compressed mode PHR symbols have the same energy as its PSDUsymbols. Comparing the compressed data rate performance with SECDEDencoded PHR versus BCH(15, 7) encoded PHR, the impact of using BCH(15,7)code can be clearly seen-it improves performance by roughly 0.3 dB forCFOs of 0 ppm, and 20 ppm. Other, stronger binary block codes shouldalso be considered, e.g., the BCH(31,11) code.

We also studied using ⅛ convolutional code with Hamming free distance of21 (see, J. Proakis, Digital Communications, ser. Electrical engineeringseries. McGraw-Hill, 2001, p. 495). We discovered that this code couldbe generated via the current encoder shown in FIG. 14. Instead of Eq. 1,this code uses the following spreading sequence depending on the encodedbit g₀ ^((k)):

$\begin{matrix}{s^{(k)} = \left\{ \begin{matrix}{{s_{0} = \left\lbrack {- {,{- {,{- {,{- {,{- {,{- {,{- {, -}}}}}}}}}}}}}} \right\rbrack},\ {g_{0}^{(k)} = 0},} \\{{s_{1} = \left\lbrack {+ {,\ {+ {,{- {,{- {,{- {,{+ {,\ {+ {,\  +}}}}}}}}}}}}}} \right\rbrack},\ {g_{0}^{(k)} = 1}}\end{matrix} \right.} & {{Eq}.\mspace{14mu} 4}\end{matrix}$

while dependence on the bit g₁ ^((k)) is the same as in Eq. 2. Noticethat the sequences s₀ and s₁ are not orthogonal. The squared Euclidianfree distance of this code equals 84, versus 80 for any code that usestwo orthogonal sequences. Hence, theoretically, the coding gainimprovement of using this code on a AWGN channel is:

10 log 10(84/80)=0.21 dB  Eq. 5:

Since the code can be produced via the existing convolutional encoder,it can be also optimally decoded by the existing Viterbi decoder. Theonly thing that is changed is the way the Viterbi metrics arecalculated. As in Eq. 3, descrambled chips are correlated with the twosequences: s₀ and s₁ to produce equivalent Viterbi metrics. The belowMatlab code snippet shows this:

if this.proakisCode this.despreadOntime0 = dscrmbld * [−1 −1 −1 −1 −1 −1−1 −1]′; this.despreadOntime1 = dscrmbld * [ 1  1 −1 −1 −1  1  1  1]′;elseIn our current Matlab testbench, there is a single shared xml control,which switches between using the orthogonal code described above andthis code for the compressed data mode; it is shown below with itsdefault value:

<proakisCode> false </proakisCode>

As shown in FIG. 10 of our Second Parent Provisional, the 10⁻² PERperformance of this code on AWGN channel is roughly 0.25 dB better thanthe performance of the orthogonal code. This agrees with our theoreticalprediction. However, as shown in FIG. 11 of our Second ParentProvisional, the 10⁻² PER performance on IEEE CM1 channel is about 0.5dB worse than the performance of the orthogonal code.

The results set forth in Sec. 4.2 of our Second Parent Provisional wasbased on the IEEE CM1 model then implemented in our trunk testbench.This model, however, was not completely implemented as the channel modeldocument prescribes. Namely, phases of the paths were set to all zero(0), instead of random. Furthermore, frequency selectivity of thechannel, represented by the K parameter in the channel model document,was not implemented. For this reason, we implemented a new channel modelimplementation which included both of these effects. Performancecomparison of two codes on such IEEE CMs are set forth in the SecondParent Provisional.

In summary, our simulation studies suggested that implementing our 27Mbps CMS results in a relatively small performance loss with respect tothe standard 6.8 Mbps scheme. Further, sensitivity loss was observed tobe due mostly to the 27 Mbps PHR reception error. This may be at leastpartly alleviated by using a stronger block code for the PHR errorcorrection—at this point we recommend considering the BCH(15,7) code.However, we expect the improvement to be relatively insignificant.

However, when we consider the simulation studies as a whole, we mustconclude from the relative performance of two possible codes-“Orthogonalcode” and “Proakis code”—that there is no clear winner. On one hand, the“Proakis code” does increased sensitivity on AWGN, as predicted by thetheory. On the other hand, the “Orthogonal code” appears to us to workbetter on all of the IEEE multipath channel models we considered. Hence,we conclude that implementation complexity should be the decisive factorwhen choosing between these two codes.

Since we completed our simulation studies, we have concluded that theOrthogonal code has additional advantages over the Proakis code that canbe exploited in many embodiments. By way of example, in Slides 16 and 17of our Third Parent Provisional, we note that, using the Orthogonalcode, the mean PRF can be varied so as to optimize relative parametersof different parts of the packet. In the Standards, mean PRF was allowedto vary, but only slightly, e.g., within a few percent. This flexibilitymade it easier to design, for example, automatic gain control (“AGC”)algorithms. With similar PRF across the whole frame, gain parameterswould not change significantly because the same energy level reaches thereceiver per unit of time. One consequence of introducing variable PRFis that the AGC receiver gain algorithms must be designed to accommodatesudden changes of received power, without distorting the receivedpulses. When the change in PRF happens, even if the transmitted pulseamplitudes do not change, the average receive power will change, but thereceiver should keep the gains the same to maintain the pulse amplitude.If the receiver knows when the change in PRF will happen, it canindicate to the AGC algorithm that at a certain point in receiving thesignal it should not change the gains significantly. In someembodiments, there are multiple gain stages in the receiver strip. Insuch embodiments, the equivalent of not changing the gains significantlyis to adjust one of the strip gains in the opposite direction to anotherstrip gain.

In some embodiments, it is advantageous for the transmitter to changethe pulse amplitude for different sections of the packet. This may beassociated with a PRF change, but another reason to do it is to makedifferent sections more, or less, robust. In such a case, the receiveroften knows the difference in amplitude between the pulses in eachportion of the signal, and it can change the gain by an amount thatkeeps the amplitude constant in the receiver.

To put it another way: if we know when the PRF is going to change, wecan, before that happens, freeze the gain (or the overall strip gain).Or, if we know that the pulse amplitude is going to change, rather thanlet the AGC do it automatically, we can expressly change the gain by theknown amount

Let us consider just these examples:

-   -   1. The frame typically consists of synchronization preamble,        SFD, (optional) cipher sequence, and (optional) data payload. In        one embodiment, each of these parts may be transmitted using a        substantially different PRF. For example, the synchronization        preamble may consist of length-127 “4a” standard Ipatov codes        (“PRF64”) or length-31 Ipatov ternary codes (see our Second        Parent Provisional, Table 2 on page 6 and Table 3 on page 7)        with mean PRF of 100 MHz; or some other synchronization        sequence, having different mean PRF.    -   2. The cipher sequence, if present, may also have variable mean        PRF as described in Sec. 16.2.8.3 of our Second Parent        Provisional, where two variants are considered, using mean PRF        of either 62.4 or 121 MHz.    -   3. The data payload may also have variable mean PRF. In        accordance with the Standards, the regular “4a” data modulation        mode uses mean PRF64. However, we note that it is possible to        enhance data payload performance by increasing mean payload PRF        to PRF256 (see, Sec. 16.3.4 in our Second Parent Provisional),        or by using other possible variants having PRF100, 167 or 125        (see, Slide 11 in our Third Parent Provisional).

Apart from having PRF which differs between parts of the frame, it isalso beneficial to have different symbol lengths in different parts ofthe frame. For example here, the Ipatov-31 preamble consists of shorter248 ns symbols. In the case of compressed cipher symbols (having meanPRF121) which are transmitted continuously without any gaps therebetween, it is possible to treat and process this cipher sequence asconsisting of symbols which can have any length. For example, a cipherconsisting of 64 1024 ns symbols may be treated as 128 512 ns symbols asor 256 256 ns symbols. Similarly, data payload, which is transmittedcontinuously, may be processed as consisting of symbols having manypossible lengths. In general, the benefits of using different symbollengths may include, for example, carrier recovery and timing tracking,where generally shorter symbols may make it easier to acquire and tracksynchronization, especially with high carrier frequency offsets.

Shorter length of symbols may be beneficial in some processingalgorithms. For example, with longer symbols, at high CFO, there is avery significant phase rotation occurring between symbols. As aconsequence, typical carrier recovery algorithm could fail tosynchronize. It can be demonstrated that introducing shorter symbols,thereby reducing phase rotation per symbol, reduces this problem.Shorter symbols could also result in faster performance of otheralgorithms, which typically work on a per-symbol basis. For example,instead of 32*1016 ns, they could finish the processing in 32*248 nswith the shorter Ipatov-31. This would result in shorter frames, andsavings of both transmitted energy and of the energy used for receiverprocessing.

However, we submit that using higher PRF across the whole frame would besub-optimal. For example, during transmission of the initialsynchronization preamble, there is no need to significantly increaseenergy density (related to higher PRF), because normal systemsensitivity during preamble detection and synchronization is sufficient,and generally higher than that of data payload demodulation. In otherwords, it makes more sense to us to increase PRF during the packet“bottlenecks”. For example, one such “bottleneck” is data demodulation.Therefore, using higher PRF during data transmission allows forequalizing performance of all parts of the packet by increasing therange at which the data can be successfully demodulated.

We further submit that using higher PRFs is particularly useful duringhighest bit-rate modes, where, generally, transmission of each bit takesless time. For example, our 27 Mbit/s scheme transmits each bit in 4×shorter time than the 6.8 Mbit/s scheme. If constant PRF was used, our27 Mbit/s scheme would be able to transmit 4× less energy per bit than6.8 Mbit/s scheme, which would lead to significant range reduction.Increasing PRF in this case helps recover the lost performance.

As noted above, there are significant benefits of using higher PRF inthe data payload portion of the packet. One of them intransmitter/receiver battery saving due to shorter airtime. However, insome applications, such as file transfer or video-streaming, higherbit-rates and much larger data payloads are required. In suchapplications, frames would again become very long, and, especially withincreased PRF, the Tx would be required to reduce power-per-pulse-per-1ms as per regulatory limitations. As a consequence, very long dataframes (or long ciphers) would have much shorter range.

In order to keep packets shorter (for battery saving) and still havesatisfactory range, we propose to transmit packets dis-continuously. Wesubmit that transmitting packets dis-continuously is of significantbenefit because:

-   -   1. It will help avoid the transmit power limit which is set        forth in the Standards to a fixed Tx power maximum per 1 ms of        transmission time. However, if a packet could be split into        parts and transmitted over several milliseconds, we could use        multiple Tx power limits.    -   2. Dis-continuous or bursty transmission allows for shutting        down the receiver to save power between successive transmission        bursts. To us, it makes more sense to send the packet components        using short bursts separated by silence gaps, rather than to        stretch, i.e., slow down or spread by using lower PRF,        transmission of the packet over time. The reason is that the        receiver can be shut down during the gaps, thus saving power,        which is especially important in battery-powered receivers. So,        in the example illustrated in FIG. 3, the transmitter can        operate continuously for, say, 100 μs, then remain silent for        900 μs, then resume continuous operation for 100 μs, then return        to silence for 900 μs, etc. In some embodiments, the bursts can        be concentrated, as shown in FIG. 4.    -   3. Even in cases where maximum Tx power is not an issue,        dis-continuous packets can still be very useful, for example, in        products powered by a small battery supported by a capacitor        (the capacitor is often needed because the small battery itself        cannot supply sufficiently high peak current for the UWB        circuits). Such applications include, for example, small car key        fobs or other small wearable items. In such embodiments, due to        space limitations, the size of the capacitor itself is limited.        Since the combination of small battery and small capacitor can        only supply high current for a very limited amount of time (to        power the UWB circuits), our bursty packet structures will allow        some time to recharge the capacitor before the next burst.

Both of the packet structures depicted in FIG. 3 and FIG. 4 use shortIpatov preamble symbols in order to facilitate coherent preambledetection on the expense of wrap-around of the longer channels' impulseresponses. A second common characteristic is that the energy of thepacket is concentrated in “active” periods with high-energy pulses,whereas “silent” periods are intended for the Tx and Rx to turn off asmany blocks as possible, especially analog ones, and thus save power.Thus, while the packet structure depicted in FIG. 3 allows for packetsof arbitrary lengths, the packet structure depicted in FIG. 4 isintended only for short packets, since increasing the packet lengthabove the prescribed maximum would produce mean power spectra withlevels above the −41.3 dBm/MHz limit set in the Standards. On the otherhand, synchronization in FIG. 3 is considerably more complex, since itconsists not only of a simple decision if the preamble is present ornot, but also finding a possible burst position and thus require longerpreamble detection times. Furthermore, Tx and especially Rx duty-cyclingin FIG. 3 is considerably less efficient, since parts of Tx and Rx needto turn on and off considerably more frequently than in FIG. 4. Also, inFIG. 3 more “on” time is required before and after every preamble burst.Considering all of the above advantages of FIG. 4 in comparison withFIG. 3, it appears to us to be rather beneficial to alleviate its oneshortcoming, i.e., packet size limitation. Let us now consider possibleapproaches to do this.

The main idea is that the concentrated packet is divided up into chunksthat are of a length that is less or equal to the maximum length of theusual concentrated packet. This maximum length produces the mean powerspectral level that meets the 41.3 dBm/MHz limit within 1 ms. Thedistance between starts of the chunks should be slightly above 1 ms. Inthis way, the 1 ms window for the power spectra density (“PSD”) wouldnever exceed above the prescribed limit.

Let us now consider possible structures of the packet, i.e., how todivide-up the packet into chunks that can be transmitted at variablePRF. It is obvious that the Ipatov preamble, start of frame delimiter(“SFD”) and physical header (“PHR”) should be transmitted together inthe same chunk. The PHR should contain a field defining the structure ofthe packet, i.e., its break-up into different chunks, which will bedependent on the needed length of different fields. Depending on thepayload length, it could also be in the same chunk with the Ipatovpreamble, SFD and PHR. The only part different from the usualconcentrated packet would be the cipher, which would now be in aseparate chunk as illustrated in FIG. 5. The beginnings of two chunksare slightly more than 1 ms apart. This structure, however, does limitthe length of the cipher to one chunk.

If longer cyphers are needed, one possible approach would be to havemultiple cipher fields. Then, as illustrated in FIG. 6, each cipherfield would be separately accumulated and processed. Note that only theexisting accumulators would be required, since the receiver facility 14would have enough time to finish before the start of the next cipher.

Packets with long payloads, e.g., for streaming applications, would havepayloads divided into chunks, as illustrated in FIG. 7. In oneembodiment, preceding each chunk a synchronization pilot could be addedfor timing and phase recovery.

Running channel impulse response analysis on different accumulators,each containing only a fraction of the total received cipher energy, isexpected to cause a degradation in performance in comparison with thecase when the whole cipher is accumulated in a single accumulator. Forthis reason, using the optional synchronization pilot in front of eachcipher, as illustrated in FIG. 8, should be considered. Then, suchcipher fields could be accumulated together in a single accumulator.

In some embodiments, mixing of the above concepts in a packet ispossible. For example, the payload may be located after the cipher, asillustrated in FIG. 9.

Applying these principles of our invention allows for efficient receiverduty cycling without reducing average transmitted power. The main openissues are the length and the structure of the pilot field. Also,related, how to do pilot timing and phase synchronization. In general,the receiver 14 should be able to estimate correct timing from thecarrier frequency offset (“CFO”) estimate within a few ns, beforereceiving the pilot field. One approach to achieving synchronization isto correlate the output of the channel match filter (“CMF”) with thepilot sequence and then periodically updating the correlation using thecarrier/timing loop.

Although we have determined that various burst patterns are possible,each has it's unique advantages and disadvantages. First, we note thatthe Tx bursts do not need to have exactly the same length or contents.For example, in FIG. 3 and FIG. 4, the 1st burst can includesynchronization preamble and data payload, and subsequent bursts cancontain parts of the cipher sequence. Or, as in FIG. 5, they can containparts of longer data payload.

The timing pattern can also be flexible. For example, slow switchingcould be implemented with each Tx burst lasting 100 μs followed by 900μs of silence, OR there could be shorter gaps, for example 50 us Tx/450μs gaps. FIG. 4 illustrates a particularly “dense” ON/OFF scheme, whereTx is ON for 1 μs followed by 7 μs silence. Preferably, the Tx patternshould be very flexible (any Tx time/any silence gap) and the duty cycleshould also be variable. In general, longer gaps make switching easierand allow for more time to recharge, for example, capacitors supportinga small battery. However, longer gaps may cause some re-synchronizationproblems, and may require additional short pilots before each burst (asshown in FIG. 7, FIG. 8, and FIG. 9). Shown in FIG. 10 is an exampletransmission of multiple data payloads but without extra SYNCs; whereas,in the example shown in FIG. 11, each chunk of the data payload ispreceded by a respective SYNC. In the example shown in FIG. 12, atransmission of a first chunk of the data payload is followed by an ACKreturn message, and then proceeds to the next chunk of the data payload;whereas, in the example shown in FIG. 13, the ACK follows transmissionof multiple chunks of the data payload. Of course, other packetstructures may be constructed, all made possible by our invention.

Although we have described our invention in the context of particularembodiments, one of ordinary skill in this art will readily realize thatmany modifications may be made in such embodiments to adapt either tospecific implementations. By way of example, it will take but littleeffort to adapt our invention for use with different communicationschemes. Further, the several elements described above may beimplemented using any of the various known semiconductor manufacturingmethodologies, and, in general, be adapted so as to be operable undereither hardware or software control or some combination thereof as isknown in this art. Alternatively, the several methods of our inventionas disclosed herein in the context of special purpose receiver apparatusmay be embodied in computer readable code on a suitable non-transitorycomputer readable medium such that when a general or special purposecomputer processor executes the computer readable code, the processorexecutes the respective method.

Thus it is apparent that we have provided several improved methods andapparatus for use in the transceiver of a wireless communication systemto transmit packets at variable PRF. We have further provided improvedmethods and apparatus to transmit packets dis-continuously. Although wehave so far disclosed our invention only in the context of apacket-based UWB communication system, we appreciate that our inventionis broadly applicable to other types of wireless communication systems,whether packed-based or otherwise, that perform channel sounding.Further, we submit that our invention provides performance generallycomparable to the best prior art techniques but more efficiently thanknown implementations of such prior art techniques.

1. A method for use in an ultra-wideband communication system fortransmitting a packet comprising first and second portions, the methodcomprising configuring a transmitter facility of the system to performthe steps of: transmitting the first portion of the packet at a selectedfirst pulse repetition frequency (“PRF”); and transmitting the secondportion of the packet at a selected second PRF different from the firstPRF.
 2. The method of claim 1: wherein the first portion of the packetcomprises at least a selected one of: a preamble; a physical header(“PHR”); a data payload; and a cipher; and wherein the second portion ofthe packet comprises at least a selected one of: the PHR; the datapayload; and the cipher.
 3. The method of claim 1 wherein thecomposition of at least the first portion of the packet is pre-defined.4. The method of claim 3: wherein the first portion of the packet isfurther characterized as comprising a synchronization sequence of aselected one of a plurality of preamble symbols; and wherein the PRF isselected as a function of the selected preamble symbol.
 5. The method ofclaim 3: wherein the first portion of the packet is furthercharacterized as comprising a synchronization sequence of a selected oneof a plurality of preamble symbols; and wherein the length of thepreamble symbol is selected to increase the carrier recovery updaterate.
 6. The method of claim 1: wherein the second portion of the packetis further characterized as comprising the data payload; and wherein thePRF is selected to reduce the transmission duration of the data payload.7. The method of claim 1: wherein the second portion of the packet isfurther characterized as comprising the data payload; and wherein thePRF is selected to increase the energy per bit of the data payload. 8.The method of claim 1: wherein the second portion of the packet isfurther characterized as comprising a selected one of a plurality ofciphers; and wherein the PRF is selected as a function of the selectedcipher.
 9. A wireless communication system configured to perform themethod of claim
 1. 10. A non-transitory computer readable mediumincluding executable instructions which, when executed in a processingsystem, causes the processing system to perform the steps of a methodaccording to claim 1.